1. Field of the Invention
The present invention relates to wireless communication apparatuses that adopt an orthogonal frequency division multiplexing (OFDM) modulation method and that perform ultra-wideband (UWB) communications in which transmission signals are spread over a wide band, and more particularly, to a multiband OFDM (MB-OFDM) wireless communication apparatus that performs UWB communications while performing frequency hopping (FP) for each OFDM symbol.
More specifically, the present invention relates to wireless communication apparatuses that perform MB-OFDM communications while avoiding, in consideration of frequency use efficiency, interference with existing communication systems that use narrow-band signals, and more particularly, to an MB-OFDM wireless communication apparatus that avoids such interference by setting an actual noise floor at an antenna terminal, not a baseband output, to a specified level or less.
2. Description of the Related Art
Recently, a wireless communication method called “ultra-wideband (UWB) communications”, which is capable of high-speed transmission of 100 Mbps or faster utilizing a very wide frequency band, has drawn attention. For example, in the United States, a UWB spectrum mask is defined by the Federal Communications Commission (FCC), and UWB transmission in the 3.1 GHz to 10.6 GHz frequency band is permitted for indoor communications. UWB communications adopts a wireless communication method for short-distance communication due to low transmission power, and is capable of high-speed wireless transmission. Thus, a personal area network (PAN) having a communication range of about 10 meters is assumed for UWB communications. Therefore, practical use of UWB communications as a wireless communication system implementing short-distance ultra-high-speed transmission has been expected.
In addition, as a technology for avoiding deterioration in transmission quality due to fading of wireless signals and for achieving higher-speed and higher-quality wireless transmission, an OFDM transmission method has been expected. In a conference for the standardization of IEEE 802.15.3a, a direct sequence spread spectrum (DSSS)-UWB method in which the spreading speed of direct-spread (DS) information signals is maximized and an OFDM-UWB method in which an OFDM modulation method is adopted were defined as UWB transmission methods. Trial production employing such methods has been conducted.
In addition, an FH method in which a used frequency band is flexibly changed has been known. In the FH method, packets are transmitted and received while the frequency is changed at random. Although communication may be interrupted due to the influence of a different system, communication is hardly ever interrupted since the frequency is continuously changed. That is, according to the FH method, coexistence with a different system can be achieved, an excellent fading resistance can be realized, and the scalability can be easily enhanced.
In the IEEE 802.15.3 standardization conference, for each of the DSSS-UWB method and the OFDM-UWB method, a multiband method (hereinafter, referred to as an “MB-OFDM method”) in which the 3.1 GHz to 10.6 GHz frequency band defined by the FCC is divided into a plurality of subbands each having a width of 528 MHz and frequency hopping is performed between the subbands is discussed.
FIG. 11 shows an example of frequency allocation defined by an MB-OFDM communication method (see, for example, “MBOFDM PHY Specification Final Release 1.0”, WiMedia Alliance, Apr. 27, 2005). In the example shown in FIG. 11, a frequency band of 5 GHz used for a wireless local-area network (LAN) is defined as a null band, and the remaining frequency band is divided into thirteen subbands. The subbands are grouped into four groups, groups A to D, and frequency is controlled for each group so that communication can be performed. The group A is a mandatory band group defined in the standard specification.
FIG. 12 shows a state in which data is transmitted while frequency hopping is performed with respect to an OFDM symbol in the time axis in the MB-OFDM method. In the example shown in FIG. 12, the group A constituted by bands #1 to #3 shown in FIG. 11 is used, frequency hopping is performed while the center frequency is changed for each OFDM symbol, and OFDM modulation adopting inverse fast Fourier transform/fast Fourier transform (IFFT/FFT) formed by 128 points is performed.
In the wireless communication environment in which a plurality of communication systems coexists, a transmission signal of one communication system may act as an interfering wave for the other systems. In particular, for UWB communications, since transmission signals are spread over wide-band frequencies, the influence of interference on existing communication systems (for example, fixed microwaves, broadcast waves, radio astronomical waves, and the like) on the same frequency band that is allocated for the UWB communications has been of concern. Thus, implementation of an interference avoidance technology in which a UWB transmission wave is emitted at a specified level or less (that is, at a very weak level) in a band in which an existing communication system exists, that is, implementation of a detect and avoid (hereinafter, referred to as DAA) technology has been regarded as being necessary.
Recently, in preparation for implementation of PANs using the UWB communication system, an active international controversy as to mitigation of the level of interference to other systems, which is caused by UWB transmission waves, has arisen. In the International Telecommunications Union-Radiocommunication Sector (ITU-R), the FCC, the European Communications Commission (ECC), and the like, legislation has been discussed. For example, in the FCC, a wireless equipment rule regarding the UWB communication system has already been decided.
In Japan, the Ministry of Internal Affairs and Communications is preparing legislation for early implementation of the UWB communication system. In a workshop session held in Aug. 25, 2005, a working group of a UWB wireless system commission in the Ministry of Internal Affairs and Communications released a provisional mask scheme about spectrum emission relating to domestic UWB transmission output regulations and comments on the implementation of interference avoidance technology. According to the release, with respect to a transmission output value permitted in the 3.4 GHz to 4.8 GHz band and the 7.25 GHz to 10.25 GHz band, if interference avoidance (DAA) technology in which the effectiveness for the fourth-generation mobile communication and broadcasting (field pickup unit (FPU)) is verified is established, for an apparatus having DAA technology, the interference level is mitigated to −41.3 dBm/MHz (the same as the frequency output regulations defined by the FCC). For apparatuses not having DAA technology, the interference level must be −70 dBm/MHz or less. In addition, discussion will be directed to the issue concerning the use of UWB being limited to indoor communications.
In a basic DAA method, the presence of a transmission signal of a different system in a UWB transmission band is examined. If a transmission signal of a different system exists in the UWB transmission band, a UWB transmission wave is emitted at a specified level or less (that is, at a very weak level). However, since a signal output from an existing communication system is a narrow-band signal, only part of the band used by the UWB communication system has an influence, if the UWB emission level is reduced to the specified level or less throughout the band by detecting the signal output from the different system, satisfactory frequency use efficiency is not achieved. Thus, it is considered that, in terms of frequency use efficiency, selectively reducing the UWB emission level to the specified level or less only in a frequency band region in which a signal of a different system is detected is desirable as a high-speed communication method.
For example, when a narrow-band carrier within a UWB band is detected, a notch is provided only in a frequency band region in which a transmission signal of a different system is detected (see FIG. 13). Thus, UWB communication using frequency band regions other than the frequency band region in which the transmission signal of the different system is detected is available while interference with the different system is avoided.
However, it is difficult for a transceiver adopting the UWB communication system corresponding to the known FCC wireless equipment rule to have the DAA function.
In a DSSS-UWB transceiver, when the presence of a different narrow-band wireless system in a used frequency band is detected, if, in order to avoid interference, a notch is provided only in a band region in which the different narrow-band wireless system is detected, a transmission waveform is distorted, resulting in failure to perform communication as a system.
In contrast, in an MB-OFDM-UWB transceiver, the FFT has a frequency detection operation. Thus, by executing analog-to-digital (A/D) conversion and FFT on a reception signal, a frequency band region in which an interfering wave exists can be examined for each subcarrier (see, for example, Japanese Unexamined Patent Application Publication No. 2004-188035, Paragraphs 0018 to 0019). If spectrum shaping is performed on a transmission signal and a notch is provided only in a subcarrier in a frequency band region in which a transmission signal of a different system is detected, it is possible to perform UWB communication using frequency band regions other than the frequency band region in which the transmission signal of the different system is detected while avoiding interference (see, for example, http://wimedia.org/en/index.asp). For example, with an active interference cancellation (AIC) technology, a saturation problem of a notch level due to interference of each subcarrier of an OFDM signal can be solved, and a notch level of 30 dB or more can be achieved (see, for example, Hirohisa Yamaguchi, “Active Interference Cancellation Technique for MB-OFDM Cognitive Radio”).
For a baseband output, the above-described methods can be realized. However, in order to actually reduce the noise floor level of an antenna terminal to −70 dBM/MHz or less, a problem in a radio frequency (RF) circuit module is more crucial. This problem will be considered for a receiver and a transmitter, individually.
FIG. 14 shows an example of a configuration of a receiver used in an MB-OFDM system (see, for example, Anuj Batra, “03267r1P802-15_TG3a-Multi-band-OFDM-CFP-Presentation.ppt”, pp.17, July 2003). The receiver shown in FIG. 14 adopts a Zero-IF configuration. In the Zero-IF method, an intermediate frequency (IF) stage is deleted. After amplifying a signal received by an antenna, the receiver performs direct frequency conversion on a baseband signal by multiplying the amplified signal by a local frequency fLO. In the example shown in FIG. 14, local (LO) signals cos(2πfLO) and sin(2πfLO) having a frequency the same as the center frequency of an RF signal are used for frequency conversion of a reception signal in an I-axis and a Q-axis. After the frequency conversion is executed, a lower frequency band is extracted through a low-pass filter (LPF), amplification is performed by a variable gain amplifier (VGA), and A/D conversion is performed. Then, a signal in a time axis is converted into a signal in a frequency axis by FFT, demodulation is performed on each carrier, and information sent as an original serial signal is reproduced. In the Zero-IF receiver, for example, when the bands of the group A shown in FIG. 11 are used, three frequencies, 3432 MHz, 3960 MHz, and 4488 MHz, which are the same as the center frequencies of RF signals, are used as local signals LO.
In the Zero-IF method, since no IF filter is used, a receiver having a wider bandwidth can be easily achieved, thus increasing the flexibility of the configuration of the receiver. On the contrary, since a reception frequency is the same as a local frequency, due to the local signal itself, a DC component, that is, a DC offset, is generated in a down-converter output (see, for example, Asad A. Abidi, “Direct-Conversion Radio Transceivers for Digital Communications” (IEEEJ. Solid-State Circuits, vol. 30, no. 12, pp. 1399-1410, 1995). Here, the position of a baseband signal of 0 Hz in the OFDM modulation method is referred to as “DC”.
Self-mixing of a local signal is generated, as shown in FIG. 15, when part of the local signal leaked from the receiver main unit toward the antenna is reflected at the antenna and returned to the receiver main unit and the returned part is multiplied by the local signal in a mixer. Alternatively, after part of the local signal is emitted toward the outside through the antenna, a reflected wave may be received at the antenna and mixed with the local signal. For example, the amplitude of the local signal is 0.5 V, the total gain of a low-noise amplifier (LNA) and the mixer is 30 dB, and −70 dB attenuation is achieved from reflection of the leakage of the local signal at the antenna to return to a point A shown in FIG. 15. In this case, the DC offset of an output of the mixer is 2.5 mV. In contrast, since the signal level of a desired wave is at least about −74 dBm, an output of the mixer exhibits −44 dBm=1.4 mVrms. Accordingly, the DC offset is greater than the signal level of the desired wave.
In the MB-OFDM-UWB system, since frequency hopping is performed for each OFDM symbol (see FIG. 12 and the above descriptions), the frequency of the local signal changes for each frequency hopping operation. In such a wide band, the reflection coefficient of the antenna is different according to the frequency of a local signal. Thus, the DC offset in the down-converter output generated by self-mixing changes in accordance with frequency hopping.
The frequency of frequency hopping is 3.2 MHz, which is the same as the symbol rate. Thus, the DC offset is changed in steps at a period of 1/3.2 MHz=312.5 nanoseconds, as shown in FIG. 16. When DC offset components, which are generated in steps, are viewed in terms of a frequency range after execution of FFT, interference occurs with respect to a baseband desired signal, as shown in FIG. 17.
In a normal receiver configuration, a VGA is disposed downstream of a down-converter output, and the gain of the VGA is controlled from a baseband signal processor side such that a reception signal has an optimal dynamic range (that is, maintains a target level) in an A/D converter, which is disposed in the downstream. Since a very large gain is obtained in the VGA, even if a DC offset component generated in the down-converter output is very small, a very large DC offset component is obtained as a VGA output. For example, if an RF circuit module is designed and produced as an RF complementary metal-oxide semiconductor (CMOS) circuit based on a wiring rule of 0.13 micrometers, the power-supply voltage is 1.2 V. Thus, in a circuit configuration in which MOS transistors are stacked vertically, a large DC offset component collapses the drain-source voltage Vds. Thus, a desired characteristic is not achieved.
In order to delete a DC offset and solve the above-described problems, generally, capacitors are inserted in series between stages of mixer outputs (see FIG. 18). In this case, a capacitor C and a circuit impedance R constitute a first-order high-pass filter (HPF). The cut-off frequency of a frequency response is 1/(2πCR), and the convergence time of a step response is 2πCR.
However, inserting the HPF prevents a baseband unit of the receiver from obtaining information on a frequency in the vicinity of the local frequency of the receiver. Thus, in the vicinity of the local frequency band, a signal from a different system is not detected. That is, DAA is not achieved.
In addition, in a normal UWB transmitter, a double-balanced mixer is used as an orthogonal modulator (MOD). If a mixer of this type performs an ideal operation, a differential signal is cancelled out between RF and baseband and between LO and RF. Thus, feed through does not exist. However, in the actual IC, due to asymmetry caused by relative variation of elements and no execution of ideal square-wave switching, feed through exists between RF and baseband and between LO and RF. Thus, a carrier leakage is generated in an MOD output.
Generation of a carrier leakage in an MOD output will be described with reference to FIGS. 19A to 19C and FIG. 20. Orthogonal modulation uses local signals having a phase difference of 90 degrees. The local signals of the phase difference of 90 degrees are multiplied with an I-axis signal having the same phase channel and a Q-axis signal being orthogonal to the I-axis signal. This multiplication circuit ideally operates so as to suppress the local signals. Actually, however, a carrier leakage is generated due to feed through caused by imbalance of circuit elements. In addition, when deviation of DC bias exists in the I-axis and the Q-axis, since the multiplication circuit does not suppress the local signals, a carrier leakage is also generated.
A local signal can be mathematically expressed as a Fourier series represented by equation (1). Here, the conversion gain of MOD_MIX is assumed as being 0 dB. In addition, the operation of the double-balanced MOD_MIX is regarded as being ideal square-wave switching.
                              LO          ⁡                      (            t            )                          =                              1            2                    -                                    2              π                        ⁢                                          ∑                                  n                  =                  1                                ∞                            ⁢                                                          ⁢                                                                    (                                          -                      1                                        )                                    n                                ·                                                                            cos                      ⁡                                              (                                                                              2                            ⁢                            n                                                    -                          1                                                )                                                              ⁢                                          ω                      LO                                        ⁢                    t                                                                              2                      ⁢                      n                                        -                    1                                                                                                          (        1        )            
A bias component of a signal I of I and Q differential signals and a bias component of an inversion signal IX (or a signal Q and an inversion signal QX) are represented by A1 and A2, respectively. Since an inversion signal LOX of a local signal is represented by 1-LO(t), a signal component of a double-balanced MOD_MIX output is represented by equation (2).
                                                                                          S                  out                                ⁡                                  (                  t                  )                                            =                            ⁢                                                                                          S                      in_p                                        ⁡                                          (                      t                      )                                                        ⁡                                      [                                                                                            L                          _                                                ⁢                                                                              O                            _                                                    ⁡                                                      (                            t                            )                                                                                              -                                              LO                        ⁡                                                  (                          t                          )                                                                                      ]                                                  -                                                                            S                      in_n                                        ⁡                                          (                      t                      )                                                        ⁡                                      [                                                                                            L                          _                                                ⁢                                                                              O                            _                                                    ⁡                                                      (                            t                            )                                                                                              -                                              LO                        ⁡                                                  (                          t                          )                                                                                      ]                                                                                                                          =                            ⁢                                                                    4                    π                                    ⁢                                      (                                                                  A                        1                                            -                                              A                        2                                                              )                                    ⁢                  cos                  ⁢                                                                          ⁢                                                            ω                      LO                                        ⁡                                          (                      t                      )                                                                      +                                                      2                    π                                    ⁢                                                            S                      1                                        ·                                                                                                                                        ⁢                                                [                                                                                    cos                        ⁡                                                  (                                                                                    ω                              LO                                                        -                                                          ω                              IN                                                                                )                                                                    ⁢                      t                                        +                                                                  cos                        ⁡                                                  (                                                                                    ω                              LO                                                        +                                                          ω                              IN                                                                                )                                                                    ⁢                      t                                                        ]                                +                …                                                                        (        2        )            
That is, if the bias component A1 of the signal I of the I and Q differential signals is different from the bias component A2 of the inversion signal IX, feed through of the local signal occurs. If it is assumed that the conversion gain of MOD_MIX is 0 dB as described above, calculation based on the assumption shows that a carrier leakage of −67 dBm is generated at MOD_MIX output even if the difference in the bias component of the I signal and the bias component of the IX signal is 0.1 mV.
WiMedia Alliance has decided upon the use of the UWB wireless system for a physical layer of a wireless universal serial bus (USB). In this case, in terms of the cost of a UWB device, it is difficult to adjust bias components of the I and IX signals to 0.1 mV or less in a production line for each USB device.
As described above, in the receiver, since a dead band region in which no signal is detected is generated in the vicinity of a local frequency of the receiver, DAA is not achieved. Thus, in accordance with the provisional mask scheme described above released by the Ministry of Internal Affairs and Communications, it is necessary to reduce the output level of a transmission signal to −70 dBm/MHz or less. In the transmitter, since a carrier leakage is generated by feed through of a local signal caused by an error in the amplitudes and phases of I and Q differential signals, it is very difficult to satisfy the above-mentioned requirement.
In addition, in order to solve the problem relating to the DC offset in the receiver having the Zero-IF configuration, a receiver having a Low-IF configuration using a lower IF is known. In general, unification of architectures of a transmitter and a receiver is basic to the design of communication apparatuses. However, in a transmitter having the Low-IF configuration, there is a problem of an image spurious component caused by IQ imbalance when orthogonal modulation is performed (an image spurious component in an orthogonal modulator is generated only in the Low-IF configuration, and an image spurious component is not generated in the Zero-IF configuration). If a detection signal exists above a local frequency, as shown in FIG. 21A, since an image frequency is generated below the local frequency, it is necessary not only to avoid an upper subband but also to avoid the image frequency in a subcarrier.
For example, if there is a gain error of 5% between an I-axis signal and a Q-axis signal, an image rejection of −32 dBc is achieved, as represented by equation (3).Image rejection=20 log(2.5/100)=−32.0[dBc]  (3)
In addition, if there is a phase error of 5 degrees between the I-axis signal and the Q-axis signal, an image rejection of −27.2 dBc is achieved, as represented by equation (4).Image rejection=20 log(tan(π·2.5/180))=−27.2[dBc]  (4)
In addition, if there is a gain error of 5% and a phase error of 5 degrees between the I-axis signal and the Q-axis signal, an image rejection of −21.7 dBc is achieved, as represented by equation (5).
                    |                                                                              Image                  ⁢                                                                          ⁢                  rejection                                =                                  10                  ⁢                                                                          ⁢                                      log                    ⁡                                          (                                                                                                    (                                                          2.5                              /                              100                                                        )                                                    2                                                +                                                                              tan                            ⁡                                                          (                                                              π                                ·                                                                  2.5                                  /                                  180                                                                                            )                                                                                2                                                                    )                                                                                                                                              =                                  -                                      21.7                    ⁡                                          [                                              dB                        ⁢                                                                                                  ⁢                        c                                            ]                                                                                                                              (        5        )            
As described above, due to a spurious image component caused by IQ imbalance when orthogonal modulation is performed, two types of interference avoidance technology, subband avoidance and subcarrier avoidance, are necessary at the same time. Thus, there is an increase in the burden regarding the adoption of countermeasures for DAA in a baseband unit.